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LTC1538CG-AUX

Part # LTC1538CG-AUX
Description IC REG CTRLR BUCK PWM CM 28-SSOP
Category IC
Availability In Stock
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Linear Technology
Date Code: 9825
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Technical Document


DISCLAIMER: The information provided herein is solely for informational purposes. Customers must be aware of the suitability of this product for their application, and consider that variable factors such as Manufacturer, Product Category, Date Codes, Pictures and Descriptions may differ from available inventory.

19
LTC1538-AUX/LTC1539
APPLICATIONS INFORMATION
WUU
U
stable input to the voltage-controlled oscillator. The filter
components C
LP
and R
LP
determine how fast the loop
acquires lock. Typically, R
LP
= 10k and C
LP
is 0.01µF to 0.1µF.
The low side of the filter needs to be connected to SGND.
The PLL LPF pin can be driven with external logic to obtain
a 1:1.9 frequency shift. The circuit shown in Figure 9 will
provide a frequency shift from f
O
to 1.9f
O
as the voltage on
V
PLLLPF
increases from 0V to 2.4V. Do not exceed 2.4V on
V
PLLLPF
.
SFB1 Pin Operation
When the SFB1 pin drops below its ground referenced
1.19V threshold, continuous mode operation is forced. In
continuous mode, the large N-channel main and synchro-
nous switches are used regardless of the load on the main
output.
In addition to providing a logic input to force continuous
synchronous operation, the SFB1 pin provides a means to
regulate a flyback winding output. The use of a synchro-
nous switch removes the requirement that power must be
drawn from the inductor primary in order to extract power
from the auxiliary winding. With the loop in continuous
mode, the auxiliary output may be loaded without regard
to the primary output load. The SFB1 pin provides a way
to force continuous synchronous operation as needed by
the flyback winding.
The secondary output voltage is set by the turns ratio of the
transformer in conjunction with a pair of external resistors
returned to the SFB1 pin as shown in Figure 4a. The
secondary regulated voltage V
SEC
in Figure 4a is given by:
VNV V
R
R
SEC OUT
≈+
()
>+
1 1 19 1
6
5
.
where N is the turns ratio of the transformer, and V
OUT
is
the main output voltage sensed by SENSE
1.
Auxiliary Regulator/Comparator
The auxiliary regulator/comparator can be used as a
comparator or low dropout regulator (by adding an exter-
nal PNP pass device).
When the voltage present at the AUXON pin is greater than
1.19V the regulator/comparator is on. The amplifier is
stable when operating as a low dropout regulator. This
same amplifier can be used as a comparator whose
inverting input is tied to the 1.19V reference.
The AUXDR pin is internally connected to an open drain
MOSFET which can sink up to 10mA. The voltage on
AUXDR determines whether or not an internal 12V resis-
tive divider is connected to AUXFB as described below. A
pull-up resistor is required on AUXDR and the voltage
must not exceed 28V.
Figure 9. Directly Driving PLL LPF Pin
18k
3.3V OR 5V
PLL LPF
2.4V
MAX
LTC1538 • F09
Low Battery Comparator
The LTC1539 has an on-chip low battery comparator
which can be used to sense a low battery condition when
implemented as shown in Figure 10. This comparator is
active during shutdown allowing battery charge level
interrogation prior to and after powering up part or all of
the system. The resistor divider R3/R4 sets the compara-
tor trip point as follows:
VV
R
R
LBITRIP
=+
119 1
4
3
.
The divided down voltage at the negative (–) input to the
comparator is compared to an internal 1.19V reference. A
20mV hysteresis is built in to assure rapid switching. The
output is an open drain MOSFET and requires a pull-up
resistor. This comparator is active when both the RUN/
SS1 and RUN/SS2 pins are low. The low side of the resistive
divider needs to be connected to SGND.
Figure 10. Low Battery Comparator
+
LBI
V
IN
SGND
LBO
R4
R3
1538 F10
1.19V REFERENCE
LTC1539
20
LTC1538-AUX/LTC1539
APPLICATIONS INFORMATION
WUU
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With the addition of an external PNP pass device, a linear
regulator capable of supplying up to 0.5A is created. As
shown in Figure 11a, the base of the external PNP con-
nects to the AUXDR pin together with a pull-up resistor.
The output voltage V
OAUX
at the collector of the external
PNP is sensed by the AUXFB pin.
The input voltage to the auxiliary regulator can be taken
from a secondary winding on the primary inductor as
shown in Figure 11a. In this application, the SFB1 pin
regulates the input voltage to the PNP regulator (see SFB1
Pin Operation) and should be set to approximately 1V to
2V above the required output voltage of the auxiliary
regulator. A Zener clamp diode may be required to keep the
secondary winding resultant output voltage under the 28V
AUXDR pin specification when the primary is heavily
loaded and the secondary is not.
The AUXFB pin is the feedback point of the regulator. An
internal resistor divider is available to provide a 12V output
by simply connecting AUXFB directly to the collector of the
external PNP. The internal resistive divider is switched in
when the voltage at AUXFB goes above 9.5V with 1V built-
in hysteresis. For other output voltages, an external resis-
tive divider is fed back to AUXFB as shown in Figure 11b.
The output voltage V
OAUX
is set as follows:
VV
R
R
OAUX
=+
<
=≥
119 1
8
7
. 8V AUX DR < 8.5V
V 12V AUX DR 12V
OAUX
When used as a voltage comparator as shown in Figure
11c, the auxiliary block has a noninverting characteristic.
When AUXFB drops below 1.19V, the AUXDR pin will be
pulled low. A minimum current of 5µA is required to pull up
the AUXDR pin to 5V when used as a comparator output in
order to counteract a 1.5µA internal pull-down current source.
Efficiency Considerations
The efficiency of a switching regulator is equal to the
output power divided by the input power times 100%. It is
often useful to analyze individual losses to determine what
is limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as:
Efficiency = 100% – (L1 + L2 + L3 + ...)
AUXDR
AUXFBSFB1
AUXON
+
+
1538 F11b
V
SEC
SECONDARY
WINDING
1:N
ON/OFF
V
OAUX
R6
10µF
R5
R8
R7
R6
R5
V
SEC
= 1.19V > (V
OAUX
+ 1V)1 +
()
LTC1538-AUX/
LTC1539
Figure 11b. 5V Output Auxiliary Regulator Using
External Feedback Resistors
Figure 11a. 12V Output Auxiliary Regulator
Using Internal Feedback Resistors
LTC1538-AUX/
LTC1539
AUXDR
AUXFBSFB1
AUXON
+
+
1538 F11a
V
SEC
SECONDARY
WINDING
1:N
ON/OFF
V
OAUX
12V
R6
10µF
R5
R6
R5
V
SEC
= 1.19V > 13V1 +
()
Figure 11c. Auxiliary Comparator Configuration
+
AUXON
AUXFB
ON/OFF
INPUT
V
PULL-UP
< 7.5V
AUXDR
OUTPUT
1538 F11c
1.19V REFERENCE
LTC1538-AUX/LTC1539
21
LTC1538-AUX/LTC1539
APPLICATIONS INFORMATION
WUU
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the resistance of one MOSFET can simply be summed
with the resistances of L and R
SENSE
to obtain I
2
R
losses. For example, if each R
DS(ON)
= 0.05, R
L
=
0.15 and R
SENSE
= 0.05, then the total resistance is
0.25. This results in losses ranging from 3% to 10%
as the output current increases from 0.5A to 2A. I
2
R
losses cause the efficiency to roll off at high output
currents.
4. Transition losses apply only to the topside MOSFET(s)
and only when operating at high input voltages (typically
20V or greater). Transition losses can be estimated from:
Transition Loss 2.5(V
IN
)
1.85
(I
MAX
)(C
RSS
)(f)
Other losses including C
IN
and C
OUT
ESR dissipative
losses, Schottky conduction losses during dead-time,
and inductor core losses, generally account for less
than 2% total additional loss.
Checking Transient Response
The regulator loop response can be checked by looking at
the load transient response. Switching regulators take
several cycles to respond to a step in DC (resistive) load
current. When a load step occurs, V
OUT
shifts by an
amount equal to (I
LOAD
)(ESR) where ESR is the effective
series resistance of C
OUT
. I
LOAD
also begins to charge or
discharge C
OUT
generating the feedback error signal which
forces the regulator loop to adapt to the current change
and return V
OUT
to its steady-state value. During this
recovery time V
OUT
can be monitored for overshoot or
ringing which would indicate a stability problem. The I
TH
external components shown in Figure 1 will prove ad-
equate compensation for most applications.
A second, more severe transient is caused by switching in
loads with large (> 1µF) supply bypass capacitors. The
discharged bypass capacitors are effectively put in parallel
with C
OUT
, causing a rapid drop in V
OUT
. No regulator can
deliver enough current to prevent this problem if the load
switch resistance is low and it is driven quickly. The only
solution is to limit the rise time of the switch drive so that
the load rise time is limited to approximately (25)(C
LOAD
).
Thus a 10µF capacitor would require a 250µs rise time,
limiting the charging current to about 200mA.
where L1, L2, etc. are the individual losses as a percentage
of input power.
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC1538-AUX/LTC1539 circuits. LTC1538-AUX/
LTC1539 V
IN
current, INTV
CC
current, I
2
R losses and
topside MOSFET transition losses.
1. The V
IN
current is the DC supply current given in the
Electrical Characteristics which excludes MOSFET driver
and control currents. V
IN
current typically results in a
small (<< 1%) loss which increases with V
IN
.
2. INTV
CC
current is the sum of the MOSFET driver and
control currents. The MOSFET driver current results
from switching the gate capacitance of the power
MOSFETs. Each time a MOSFET gate is switched from
low to high to low again, a packet of charge dQ moves
from INTV
CC
to ground. The resulting dQ/dt is a current
out of INTV
CC
which is typically much larger than the
control circuit current. In continuous mode, I
GATECHG
=
f(Q
T
+ Q
B
), where Q
T
and Q
B
are the gate charges of the
topside and bottom side MOSFETs. It is for this reason
that the large topside and synchronous MOSFETs are
turned off during low current operation in favor of the
small topside MOSFET and external Schottky diode,
allowing efficient, constant-frequency operation at low
output currents.
By powering EXTV
CC
from an output-derived source,
the additional V
IN
current resulting from the driver and
control currents will be scaled by a factor of Duty Cycle/
Efficiency. For example, in a 20V to 5V application,
10mA of INTV
CC
current results in approximately 3mA
of V
IN
current. This reduces the midcurrent loss from
10% or more (if the driver was powered directly from
V
IN
) to only a few percent.
3. I
2
R losses are predicted from the DC resistances of the
MOSFET, inductor and current sense R. In continuous
mode the average output current flows through L and
R
SENSE
, but is “chopped” between the topside main
MOSFET and the synchronous MOSFET. If the two
MOSFETs have approximately the same R
DS(ON)
, then
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